The introduction by the 3rd-Generation Partnership Project (3GPP) of standards for the Long-Term Evolution (LTE) and LTE-Advanced wireless communication have stimulated further interest in the design of multi-mode, multi-band radio transceivers. The direct conversion receiver, also known as the homodyne receiver, is central to the multi-band approach to multi standards transceiver architecture.
It is well known that flicker noise of the down-conversion mixer in a homodyne (zero-IF) receiver appears in the baseband, at the signal band of interest. Interest in reduced flicker noise has been one of the driving forces behind recently renewed attention to passive mixer circuits.
In mature mixer designs, based on Gilbert-type active mixers, switching transistors driven with DC bias current steer the radio frequency (RF)-to-baseband conversion. However, flicker noise at the output of the mixer is proportional to the DC current through Switches. Another limiting factor to the performance of the Gilbert-cell mixer is the voltage-to-current conversion circuitry used in these mixers. This circuitry puts limits on the linearity of the RF-to-baseband conversion.
For these and other reasons, the passive FET (field-effect transistor) mixer topology (typically based on the metal-oxide-semiconductor FET, or MOSFET, or, somewhat more generally, based on the insulated-gate FET, or IGFET) offers potential improvements in noise (low flicker-noise corner) and better linearity response, compared to the active Gilbert-cell mixer. FIG. 1 illustrates an example of a four-phase passive mixer with in-phase (I) and quadrature (Q) outputs. Two passive mixer branches comprising four transistors each are connected in parallel and operated in quadrature. Thus, each transistor will be activated when a local oscillator (LO) signal at its gate has a sufficiently positive value, i.e., when the LO signal voltage exceeds the threshold value of the FET. Each mixer branch is connected to provide signal paths from input signal terminals RF+/RF− to first and second intermediate frequency (IF) terminals IFI and IFQ. Conventionally, the first mixer branch is driven by a first signal LOI+ and its inverse signal LOI−, having phases φ and φ+π, respectively. The second mixer branch is driven by a second LO signal LOQ+ and its inverse signal LOQ−, having phases φ+π/2 and φ+3π/2, respectively.
In operation, two LO signals in the circuit of FIG. 1 will have positive values simultaneously. Although the transistors are operated so that the IF terminals are generating the IF signals alternately, a path (short circuit) is created between the IF terminals of the two mixers when any two LOI and LOQ signals are high. For example, this is the case when LOI+ and LOQ+ have positive values simultaneously.
Theoretical noise figures for passive FET mixers can be estimated for several possible configurations. One mixer configuration, for example, is a passive two-phase mixer with a resistive In-Phase/Quadrature-Phase (I/Q) split. An example of this configuration is shown in FIG. 2. Assuming a 1:2 balun, this mixer configuration has the following minimum theoretical noise figure:F=6+(2πN/G),  (1)where N is balun turn ratio and G is the gain of the signal chain from the RF input of the mixer to the output of the first baseband operational amplifier (op-amp).
As can be seen from Equation (1), the minimum noise figure can never be lower than 6, which is 7.8 dB. Normally the gain in the receiver chain is limited. The gain is adopted by variable gain amplifier, VGA circuitry. A gain of 12.59=22 dB in the chain results in a noise figure of:F12.59,twophaseIQ=6.9982=8.4 dB.
Another possible configuration is the four-phase mixer with time-split I/Q architecture. An example of this configuration is illustrated in FIG. 3. A timing diagram for LO signals is pictured in FIG. 4. Again assuming a 1:2 balun, the following minimum theoretical noise figures can be calculated:F=2+(sqrt(2)πN/G).  (2)Given a gain of 12.59=22 dB in the receiver chain, the resulting noise factor is F12.59,fourphaseIQ=2.7=4.3 dB.
Not that the above calculations of noise figures take only thermal noise from the terminations and the first feedback resistor of the first baseband op-amp into account. Only the first upper and lower sidebands are considered. The switching devices are ideal and more noise sources will contribute to the real noise figure.
The above described architectures for resistive I and Q split and four-phase time domain I and Q split look very similar to each other but their functionalities are different. A main difference is in the LO signal, which is a two-phase LO signal in case of the resistively split architecture, versus a four-phase LO signal for the time-split I/Q architecture. Another difference is the resistor termination value at the input side of the mixer, which will be doubled in the case of the resistively split I and Q design. Since both the I and Q mixers are always “on”, the input resistor terminations are paralleled and therefore doubled in value compared to the four-phase version, resulting in a noise penalty. This noise penalty is accounted for in expression (1), above.
While existing mixers based on the four-phase passive mixer architecture offer advantages over the Gilbert-cell mixer, techniques are needed to further enhance the noise performance of these passive mixers.